This invention relates to radio communications systems, and more particularly to apparatus and methods for coupling a plurality of transmitters having non-identical but relatively close output frequencies to a single antenna.
In the recent past, radio communications systems, such as cellular telephone systems and trunked radio systems, have been developed which can provide vast amounts of capacity to handle communications traffic between mobile and portable subscribers and land-based communications systems. In some cases, these systems can support the communications needs of many thousands of users.
Such radio systems achieve their significant communications capacity, in part, by dividing the geographical area in which coverage is desired into small regions (or cells) and deploying therein land-based radio transmitting and receiving equipment sufficient to meet the traffic requirements of that region. Because each region is relatively small, both the mobile stations and the land stations may use relatively low transmitter power. As a result, when a channel is in use in a particular region, that channel may be simultaneously reused in a non-adjacent region only a short distance away.
However, another key to the large capacity of modern radio communications systems is providing a very large number of available radio channels in each small region. For example, in the domestic cellular telephone service, 416 channels are available to each system operator, and an operator may typically allocate 25-30 of those channels for use in a particular region or cell. The large number of available channels span a wide frequency range. For example, in the domestic cellular service, the channels allocated for transmitting from the base station to the subscriber terminal extend from 869 MHz to 894 MHz, a range of about 25 MHz.
For a variety of reasons, the land-based radio equipment for all of the channels provided in each region or cell is generally located at a very small number of places therein. Typically a single base station is used in each region, but in some systems remote base stations may be provided to increase capacity or to avoid coverage defects due to buildings or topography. Accordingly, a single base station site may have 30 or more pairs of radio transmitters and receivers.
A significant problem in the design of base station equipment where a large number of channels must operate simultaneously is attaching the transmitters and receivers to suitable antennas. Although it is conceivable that a separate antenna could be provided for each receiver and transmitter, that solution has many disadvantages. Conventional antennas generally must be located some distance away from other antennas for proper operation, and therefore, providing separate antennas for each transmitter and receiver would require unacceptable amounts of "real-estate" on the towers or buildings on which they are mounted. A transmission line must be provided to connect each antenna to the associated radio equipment. The size and weight of a large bundle of transmission lines is unacceptable in many installations. Further, since the antennas and the transmission lines are exposed to the environment, a large number of antennas and transmission lines create a significant maintenance burden. In addition, the antennas and transmission lines are costly.
Designers of radio systems have sought and developed ways to connect a large number of transmitters or receivers to a single antenna. Special problems occur when attempting to connect multiple transmitters to a single antenna, because the transmitters, by definition, generate radio-frequency (RF) energy. In general, it is not feasible to simply connect several transmitters, in parallel, to an antenna, because each transmitter would appear as a load to the other transmitters. Thus, the RF energy produced by one transmitter would, at least in part, be dissipated in the other transmitters. This is inefficient, because a substantial portion of each transmitter's output may be dissipated in other transmitters instead of being radiated by the antenna. In addition, the RF energy received from other transmitters is dissipated in the form of heat, so that if a transmitter receives sufficient RF energy from other transmitters, it may be damaged.
Accordingly, radio system designers have developed "combiners" for properly coupling the RF energy produced by several transmitters to a single antenna/transmission line. A simplified block diagram of a prior art combiner 100 is shown in FIG. 1. The conventional combiner 100 chosen for illustration herein is adapted for use with up to 20 transmitters, but combiners of smaller or larger capacity have been constructed. Further, although the conventional combiner 100 is of a design suitable for use in the 750-1250 MHz frequency range, combiners of various designs are available for a wide range of frequencies.
As best seen in FIG. 1, a conventional combiner comprises a plurality of input ports 110a-110t for receiving RF energy from transmitters 152 via suitable transmission lines 154. Each input port 110a-110t is connected to a respective filter means 112. Each filter means 112 is generally a relatively narrow bandpass filter having its passband centered about the frequency on which the associated transmitter 152 operates. Although the filters 112 may be implemented using a variety of technologies, the filters provided in typical commercial combiners for use in the 150-1500 MHz frequency range are implemented using cavity resonators which may include a ceramic dielectric element. The output signal from the transmitter is introduced into, and collected from, the filter 112 using any means appropriate for the type of filter being used. For the cavity resonators described herein, wire loops 138, 142 are used, but other means, such as probes, could also be used. An adjustment means 136 is provided to control the resonant frequency of the filter. The signal from the transmitter is provided at an output port 114 of the filter.
The filters 112 function to preclude RF energy produced by one transmitter from being delivered to any other transmitter. The filter passband is selected to be wide enough to pass the transmitted signal, but narrow enough to reject the frequencies on which all other transmitters at the site operate. Thus, for each transmitter, the associated filter rejects substantially all of the RF energy which may be available from the other transmitters at the site.
Although regulatory standards for modern communications systems provide for adjacent channel spacing of 15 to 50 kHz, in practice, it is difficult and expensive to construct filter elements suitable for operation of multiple transmitters on immediately adjacent channels. To partially avoid this problem, the channels selected for use at a particular site are chosen such that each operating channel is separated from adjacent operating channels by several non-operating channels.
The minimum allowed frequency difference between adjacent channels for a combiner is a design parameter and is referred to as "channel separation." Channel separation in commercial communications systems typically ranges from approximately 150 kHz to approximately 900 kHz. The channel separation is an important design parameter affecting system performance. Although a designer may accommodate the need for reduced channel separation by increasing the loaded Q of the filter, thereby increasing the slope of the filter's response curve, increasing the loaded Q also increases the insertion loss of the filter, thereby reducing the amount of transmitter-produced RF energy which is delivered to the antenna.
The remaining portions of the conventional combiner 100 are provided to achieve the physical interconnection between the output ports 114 of the filters 112 and the transmission line 132 to the antenna 134. A primary transmission line 116 connects the output port 114 of each filter to a corresponding input port 140 of one of four primary junction assemblies 118a, 118f, 118k, 118p. Each primary junction assembly 118 has five input ports 140 which are connected in parallel to a single output port 122.
A secondary transmission line 124 connects the output port 122 of each primary junction assembly 118 to a secondary junction assembly 126. The secondary junction assembly 126 has four input ports 144 which are connected in parallel to a single output port 128. The output port 128 of the secondary junction assembly 126 is connected to a transmission line 132 which is, in turn, connected to the antenna 134. Two cascaded "layers" of small junction assemblies 118 and 126 are provided instead of a single large 20-input junction assembly because it is difficult to construct large junction assemblies in which the transmission path between the input ports and the central junction has the desired transmission line characteristics.
In connecting the output ports 114 of the filters 112 to the combiner's final output port 128, it is desirable to avoid introducing reactance which may be caused by the transmission lines 116 and 124 and the junction assemblies 118 and 126. Accordingly, a conventional combiner 100 of the type described herein is typically constructed such that the effective electrical length of each transmission path from the output port 114 of a filter 112 to the final output port 128 closely approximates an integral multiple of one half wavelength at the center of the combiner's operating frequency range. A transmission line having an electrical length of exactly an integral multiple of one half wavelength is electrically "invisible" in that it contributes no reactance to the circuit.
The undesired introduction of reactance into the transmission line circuit from various combiner components significantly degrades the performance of the conventional combiner 100 of FIG. 1. Commercial antennas and transmission lines are typically designed to have input and characteristic impedances, respectively, in the range of approximately 50 to 75 ohms. An impedance mismatch between the combiner and the antenna circuit, which may be caused by undesired reactance in the combiner, causes power to be reflected back or "returned" to the transmitters. The reflected power is dissipated as heat in the transmitter and if it is sufficiently large, may damage the transmitter. In addition, any power reflected by the antenna circuit is obviously not radiated. In addition, the undesired reactance increases the insertion loss of the combiner.
The undesired reactances produced in the conventional combiner 100 of FIG. 1 are caused by two principal sources, in cooperation: the filter means 112, and the transmission line components 116, 124. The cavity resonators used to implement the filter means 112 produce little reactance themselves at exactly their resonant frequency; at frequencies far removed from their resonant frequencies, they appear as open circuits and thus also produce little reactance.
Thus, at the output frequency of a particular transmitter, there will be virtually no reactance directly contributed by the transmitter's associated cavity, which is resonant at that output frequency. However, when the cavity is connected to a length of transmission line, as it necessarily is in a combiner, the impedance of the cavity as seen through the transmission line will vary depending on the output impedance of the cavity, the characteristic impedance of the line, and the length Of the line. If the effective electrical length of the transmission line is exactly an integral multiple of one half wavelength, there will be no reactance contribution apparent from the cavity, regardless of the impedance of the transmission line.
However, if the electrical length of the transmission line is not exactly an integral multiple of one half wavelength, the transmission line will transform the impedance of the cavity, even at resonance, such that the cavity appears reactive. Typically, the transmission line lengths in a combiner will be "exactly" an integral multiple of one half wavelength only at the combiner center frequency. Thus, at all other frequencies in the operating band, errors in the effective length of the transmission line segments will produce a reactance visible at the end of the transmission line. This reactance contribution is attributed to the associated cavity.
A first reactance contribution source to be considered is a cavity far off resonance. At a frequency far from its resonant frequency, a cavity exhibits a very high resistance and appears essentially as an open circuit. From the point of view of a transmitter operating at the low end of the combiner's frequency range, transmission-lines which are intended to be one half-wavelength are "short". Thus, the open circuits presented by far-off-resonance cavities are transformed by their associated transmission lines as a pure inductance. From the point of view of a transmitter operating at the high end of the combiner's frequency range, transmission lines which are intended to be one half-wavelength are "long", and therefore, the open circuits presented by far-off-resonance cavities are transformed by their associated transmission lines as a pure capacitance.
A second reactance contribution source to be considered is a cavity at resonance. A cavity at resonance itself exhibits negligible reactance. In practical combiners, each cavity is connected to a transmission line, which is typically intended to have an electrical length of an integral multiple of one half wavelength, at the combiner center frequency. However, since most, if not all, of the transmitters and their associated cavities are tuned to a frequency other than the combiner center frequency, from the point of view of these transmitters, the transmission lines will be either "long" or short.
The amount of reactance contributed by a resonant cavity at a selected frequency, and the sign of the reactance (i.e. whether the contributed reactance is inductive or capacitive), depends on the output impedance of the cavity and whether the selected frequency is above or below the combiner center frequency.
At a frequency near the low end of the combiner operating range, the transmission line will appear short. If the output impedance of the associated cavity is higher than the characteristic impedance of the transmission line, then the "short" transmission line will transform the resistance of the cavity to an inductive reactance. At a frequency near the high end of the combiner operating range, the transmission line will appear long. If the output impedance of the associated cavity is higher than the characteristic impedance of the transmission line, then the "long" transmission line will transform the resistance of the cavity to a capacitive reactance.
Because it is generally desirable to match the combiner's output impedance to the input impedance of the antenna circuit, cavities of conventional combiners 100 have been designed with relatively high output impedances. Early combiners used cavities with output impedances in the range of 50-60 ohms. Thus, many conventional combiners behave as described above: resonant cavities contribute inductance at the low end of the operating range and capacitance at the high end of the operating range. Unfortunately, this reactance contribution operates in the same direction as the contributions from far-off-resonance cavities and associated transmission lines, and therefore, large amounts of undesired reactance contributions are produced. This results in particularly poor performance in many conventional combiners.
FIG. 7 is a Smith Chart 202 representing a computer simulation of the output match presented by a 20-channel combiner of conventional design over a range of operating frequencies at selected output impedances. The simulation calculates the output match (i.e. the match as seen at the output terminal of the combiner) as the resonant frequency of a single channel cavity is swept over the frequency range of interest. Portions of the chart above the equator 214 represent inductive reactance; portions of the chart below the equator 214 represent capacitive reactance. Curves 204, 206, 208, 210, and 212 represent the output match of the combiner using cavities having design impedances of 50, 40, 30, 20, and 10 ohms respectively. The remaining channel cavities are assumed to be substantially detuned from the frequency range of interest. Thus, the curves of FIG. 7 include the reactance contributed by the swept cavity at resonance, and the reactance contributed by the remaining cavities far from resonance.
Thus, as best illustrated by curve 204 of FIG. 7, a combiner employing a cavity having a design impedance of 50 ohms produces relatively large amounts of inductive reactance at frequencies below the combiner center frequency, and relatively large amounts of capacitive reactance at frequencies above the combiner center frequency. As noted above, large amounts of reactance contributed by combiner components degrades the performance of the combiner.
On the other hand, if the output impedance of a resonant cavity is lower than the characteristic impedance of the transmission line, then at frequencies near the low end of the combiner operating range, a "short" transmission line will transform the resistance of the cavity to a capacitive reactance. At a frequency near the high end of the combiner operating range, a long transmission line will transform the resistance of the cavity to an inductive reactance. Thus, where the cavity output impedance is low, compared to the characteristic impedance of the transmission line, the reactance contributions from resonant cavities (and associated transmission lines) are opposite in direction from the reactance contributions of far-off-resonant cavities. When this condition exists, these reactance contributions may, to some extent, compensate each other.
Accordingly, some designers of prior art combiners have sought to reduce the reactance contribution of the cavities by employing cavities having a somewhat lower design impedance. Combiners employing cavities with design impedances as low as 35 ohms have been constructed. Although the performance of such combiners may have improved somewhat due to the reduction in the reactance contributed by the cavities, the relatively low design impedance of the cavities resulted in a poor impedance match when used with a standard 50-75 ohm antenna system. In addition, the 35 ohm output impedance of the cavities did not produce sufficient reactance contributions from resonant cavities to effect the desired compensation of reactance contributions from far-off-resonant cavities. Thus, the prior-art use of low-output-impedance cavities did not result in the desired overall system performance improvement.
The computer-simulated performance of a 16-channel combiner of conventional design is summarized in FIGS. 2-3. The combiner is designed to operate in a 33 MHz bandwidth around 933.5 MHz, with a channel separation of 300 kHz. In order to illustrate worst case performance, the lowest available 16 channels within the specified operating range were selected. This condition maximizes the reactances produced by the cavities and the transmission lines. FIGS. 2a-2d present diagrams 260a-260d representing the insertion loss of a conventional combiner in the frequency range of 916 to 922 MHz. FIGS. 3a-3b present diagrams 262a-262d representing the output return loss of the conventional combiner over that frequency range.
As best seen in FIGS. 2a-2d, the insertion loss increases dramatically with frequency. At the highest selected frequency, the combiner has an insertion loss of about 4.3 dB (i.e. only about 37 percent of the original signal is available at the output of the combiner). As best seen in FIGS. 3a-3b, the output return loss decreases dramatically with frequency. At the highest selected frequency the output return loss is less than 5.0 dB (i.e. about 32 percent of the power available at the output of the combiner is returned as reflected power). Thus, a significant improvement in combiner performance is highly desirable.
In order to attempt to compensate the reactances produced by the cavities and the transmission lines, adjustable reactance elements 120 or 130 have been provided in conventional combiners at primary or secondary junctions 118, 126 respectively, or at other suitable locations. For example, in some prior art combiners, a shorted stub of adjustable length is connected in shunt to the center junction point of the primary or secondary junction assemblies.
This technique may allow a user to optimize the performance of the combiner for a particular limited range of frequencies. For example, for conventional combiners designed for operation in a particular 25 MHz band in the 900 MHz region, this reactance compensation technique may permit the user to select a 10 MHz segment in which the combiner provides acceptable performance. However, this technique has the disadvantages of limiting the usable bandwidth of the combiner and it requires manual adjustment.